Direct ac power converting apparatus

ABSTRACT

A control section controls a current-source converter in a state in which a switch is in conduction, and performs voltage doubler rectification on a voltage between a neutral phase input line and any of input lines to provide for charging of clamp capacitors. Accordingly, the clamp capacitors are charged through a resistor, which prevents an inrush current from flowing therethrough. In addition, a voltage between both ends of a pair of the clamp capacitors is higher than, for example, a voltage between both ends of a pair of capacitors. Accordingly, even if the clamp capacitors and, for example, the capacitors are electrically connected to each other in a normal operation, it is possible to prevent the inrush current from flowing from the capacitors to the clamp capacitors.

TECHNICAL FIELD

The present invention relates to a direct AC power converting apparatus,and more particularly, to a technology of preventing inrush current to acapacitor included in a direct AC power converting apparatus.

BACKGROUND ART

Non-Patent Document 1, which will be described below, discloses a directAC power converting apparatus including a clamp circuit. FIG. 24 showsthe direct AC power converting apparatus described in Non-PatentDocument 1. Note that for the sake of description of the presentinvention, reference symbols in the drawing do not necessarilycorrespond to those of Non-Patent Document 1.

It is assumed here that an IPM motor is provided on an output side ofthis direct AC power converting apparatus. When La represents aninductance per phase which corresponds to an average value of effectiveinductances of the IPM motor, i represents overload current which servesas a reference for interrupting current supply to the IPM motor, Vcrepresents voltage between both ends of a clamp capacitor, Cc representselectrostatic capacitance of the clamp capacitor, and Vs represents linevoltage of a three-phase AC power supply, and when all power stored inan inductor for three phases of the IPM motor is regenerated to theclamp capacitor, the following relational expression is satisfied.

$\begin{matrix}{\lbrack {{Expression}\mspace{14mu} 1} \rbrack \mspace{596mu}} & \; \\{{\frac{1}{2}{{La}( {i^{2} + ( \frac{i}{2} )^{2} + ( \frac{i}{2} )^{2}} )}} = {\frac{1}{2}{{Cc}( {{Vc}^{2} - ( {\sqrt{2}{Vs}} )^{2}} )}}} & (1)\end{matrix}$

Therefore, the voltage between both ends of the clamp capacitor isexpressed by the following expression.

$\begin{matrix}{\lbrack {{Expression}\mspace{14mu} 2} \rbrack \mspace{596mu}} & \; \\{{Vc} = \sqrt{{\frac{3}{2}\frac{La}{Cc}i^{2}} + {2{Vs}^{2}}}} & (2)\end{matrix}$

FIG. 25 is a graph showing the relationship between voltage between bothends and electrostatic capacitance of the clamp capacitor, which isbased on Expression (2). For example, if the power supply voltage Vs is400 V, the inductance La is 12 mH, the overload current i is 40 A, andthe electrostatic capacitance of the clamp capacitor is 10 μF, thevoltage Vc between both ends of the clamp capacitor is approximately1,800 V. The power supply value exceeds device rating 1,200 V of atransistor and a diode with power supply voltage of 400 V class.

In order to keep the voltage Vc between both ends of the clamp capacitorat approximately 750 V or lower, the electrostatic capacitance of theclamp capacitor needs to be 200 μF or larger from Expression (2) andFIG. 25.

On the other hand, inrush current at power-on increases as theelectrostatic capacitance of the clamp capacitor is increased. Here, aseries circuit in which a power supply, a reactor, a resistor and acapacitor are connected in series is taken as an example of a seriescircuit for one phase, where L represents an inductance of the reactor,R represents a resistance value of the resistor, and C representselectrostatic capacitance of the clamp capacitor. Then, a transfercharacteristic of output (current) to input (power supply voltage Vs) inthe series circuit is expressed by the following expression.

$\begin{matrix}{\lbrack {{Expression}\mspace{14mu} 3} \rbrack \mspace{596mu}} & \; \\{{G(s)} = {\frac{ic}{Vs} = {{sC}\frac{1/{LC}}{s^{2} + {{sR}/L} + {1/{LC}}}}}} & (3)\end{matrix}$

The response to step input is expressed by the following expression.

$\begin{matrix}{\lbrack {{Expression}\mspace{14mu} 4} \rbrack \mspace{596mu}} & \; \\{{G(s)} = {{{sC}\frac{1/{LC}}{s^{2} + {{sR}/L} + {1/{LC}}}\frac{1}{s}} = \frac{1/L}{s^{2} + {{sR}/L} + {1/{LC}}}}} & (4)\end{matrix}$

Here, Expression (4) is subjected to inverse Laplace transform to obtainthe response of current assuming that 1/L=D, R/L=E and 1/LC=F, then thefollowing expression is derived.

$\begin{matrix}{\lbrack {{Expression}\mspace{14mu} 5} \rbrack \mspace{596mu}} & \; \\{{{i(t)} = {\frac{D}{\omega}^{{- o}\; t}\sin \mspace{11mu} \omega \; {t\lbrack {{Expression}\mspace{14mu} 6} \rbrack}}}\mspace{554mu}} & (5) \\{{\omega = \frac{\sqrt{{4F} - E^{2}}}{2}},\mspace{14mu} {\sigma = \frac{E}{2}}} & (6)\end{matrix}$

F decreases as the electrostatic capacitance C of the capacitorincreases, and D and E remain constant irrespective of the electrostaticcapacitance C, and thus ω decreases as the electrostatic capacitance Cof the capacitor increases. Accordingly, an amplitude term D/ω excludingattenuation through time increases as the electrostatic capacitance C ofthe capacitor increases. That is, inrush current increases along with anincrease in electrostatic capacitance C of the capacitor.

When the maximum value of current is obtained assuming that a valueobtained by differentiating i(t) with respect to time is 0 (i(t)′=0)from Expression (5), the following expression is derived.

$\begin{matrix}{\lbrack {{Expression}\mspace{14mu} 7} \rbrack \mspace{596mu}} & \; \\{t = \frac{\pi - \alpha}{\omega}} & (7)\end{matrix}$

The current has the maximum value on this occasion. This maximum valueis considered to be inrush current. FIG. 26 is a graph showing therelationship between inrush current (i((π−α)/ω)) and the electrostaticcapacitance C.

As described above, the voltage between both ends of the clamp capacitorcharged with the regenerative current is approximately equal to or lowerthan 750 V, and accordingly if the electrostatic capacitance of theclamp capacitor is 200 μF, the maximum value (inrush current) of currentreaches 150 A from Expressions (6) and (7).

Patent Documents 1 to 4 disclose the technologies related to the presentinvention.

Non-Patent Document 1: Lixiang Wei and Thomas A. Lipo, “Investigation of9-switch dual-bridge matrix converter operating under low output powerfactor”, USA, IEEE, ISA 2003, vol. 1, pp. 176-181

Patent Document 1: U.S. Pat. No. 6,995,992

Patent Document 2: Japanese Patent Application Laid-Open No. 2006-54947

Patent Document 3: Japanese Patent Application Laid-Open No. 08-79963

Patent Document 4: Japanese Patent Application Laid-Open No. 02-65667

DISCLOSURE OF INVENTION Problem to be Solved by the Invention

As described above, there is a problem that inrush current to the clampcapacitor increases as electrostatic capacitance of the clamp capacitoris increased for suppressing an increase in voltage between both ends ofthe clamp capacitor due to regenerative current.

An object of the present invention is therefore to provide a direct ACpower converting apparatus capable of reducing inrush current whileincreasing electrostatic capacitance of a capacitor to prevent anincrease in voltage between both ends of the capacitor.

Means to Solve the Problem

According to a first aspect of the present invention, a direct AC powerconverting apparatus includes: a plurality of input lines (ACLr, ACLs,ACLt) to which an output of a multi-phase AC power supply (E1) includinga neutral point is applied; a positive-side DC power supply line (L1); anegative-side DC power supply line (L2) to which a potential lower thana potential applied to the positive-side DC power supply line isapplied; a current-source power converter (1) including a plurality ofswitch devices, converting a multi-phase AC voltage applied between onesof the plurality of input lines into a square-wave-shape DC voltagehaving two potentials by selection operations of the plurality of switchdevices, and supplying the DC voltage between the positive-side DC powersupply line and the negative-side DC power supply line; a plurality ofinput capacitors (Cr, Cs, Ct) each provided between the ones of theplurality of input lines and functioning as a voltage source; a firstdiode (D1) provided between the positive-side DC power supply line andthe negative-side DC power supply line and having an anode on thepositive-side DC power supply line and a cathode on the negative-side DCpower supply line side; first and second capacitors (Cc1, Cc2)connected, between the positive-side DC power supply line and thenegative-side DC power supply line, in series with the first diode; aneutral phase input line (ACLn) connecting the neutral point and a pointbetween the first capacitor and the second capacitor; a voltage-sourcepower converter (3) converting the DC voltage into a square-wave-shapeAC voltage and outputting to an inductive multi-phase load (4); aresistor (R1) inserted in any one of the plurality of input lines andthe neutral phase input line; and a control section (5) controlling theselection operations of the plurality of switch devices, performingvoltage doubler rectification on a phase voltage for one phase which isapplied between one of the plurality of input lines and the neutralphase input line, and supplying for charging of the first capacitor andthe second capacitor through the resistor.

According to a second aspect of the present invention, in the direct ACpower converting apparatus according to the first aspect, the direct ACpower converting apparatus further includes a second diode (D5)connected, between the positive-side DC power supply line (L1) and thenegative-side DC power supply line (L2), in series with the first diode(D1), and having an anode on the positive-side DC power supply line (L1)side and a cathode on the negative-side DC power supply line (L2) side,wherein the neutral phase input line (ACLn) is connected between thefirst and second diodes.

According to a third aspect of the present invention, in the direct ACpower converting apparatus according to the first or second aspect, thedirect AC power converting apparatus further includes a switch (S1)provided on the neutral phase input line (ACLn), wherein the controlsection (5) supplies for charging of the first capacitor (Cc1) and thesecond capacitor (Cc2) in a state in which the switch is in conduction,and brings the switch into nonconduction after a lapse of apredetermined period of time.

According to a fourth aspect of the direct AC power converting apparatusof the present invention, in the direct AC power converting apparatusaccording to any one of the first to third aspects, the resistor (R1) isprovided on the neutral phase input line (ACLn).

According to a fifth aspect of the direct AC power converting apparatusof the present invention, in the direct AC power converting apparatusaccording to any one of the first to third aspects, the resistor (R1) isprovided in one of the plurality of input lines (ACLr, ACLs, ACLt), thedirect AC power converting apparatus further including reactors (Lr, Ls,Lt) connected in parallel with the resistor.

According to a sixth aspect of the direct AC power converting apparatusof the present invention, in the direct AC power converting apparatusaccording to any one of the first to fifth aspects, the first capacitor(Cc1) is provided on the positive-side DC power supply line (L1) sidewith respect to the second capacitor (Cc2), and the first diode (D2) isprovided between the first capacitor and the second capacitor, thedirect AC power converting apparatus further including: a third diode(D3) having an anode connected between the first diode and the secondcapacitor and a cathode connected to the positive-side DC power supplyline; and a fourth diode (D4) having an anode connected to thenegative-side DC power supply line and a cathode connected between thefirst diode and the first capacitor.

EFFECTS OF THE INVENTION

According to the first aspect of the direct AC power convertingapparatus of the present invention, the AC voltage for one phase issubjected to voltage doubler rectification to charge the first capacitorand the second capacitor, whereby it is possible to prevent the inrushcurrent from flowing from the multi-phase AC power supply to the firstcapacitor and the second capacitor in an initial operation of thecurrent-source power converter. Since the resistor is provided in a pathof the charging, the inrush current does not flow in the charging aswell. On this occasion, the input capacitor is not electricallyconnected to the first capacitor and the second capacitor. Therefore,the inrush current does not flow from the input capacitor to the firstcapacitor and the second capacitor even if the input capacitor ischarged with voltage.

According to the second aspect of the direct AC power convertingapparatus of the present invention, in the normal operation in which thecurrent-source power converter converts a multi-phase AC voltage into aDC voltage having two potentials and the voltage-source power converterconverts the DC voltage into a square-wave-shape AC voltage, it ispossible to prevent the second capacitor from being charged/dischargedthrough the neutral phase input line, which accordingly prevents a lossof symmetry of the input currents.

According to the third aspect of the direct AC power convertingapparatus of the present invention, in the normal operation in which thecurrent-source power converter converts a multi-phase AC voltage into aDC voltage having two potentials and the voltage-source power converterconverts the DC voltage into a square-wave-shape AC voltage, connectionof the power supply to the first capacitor and the second capacitorthrough the neutral phase input line is cut off. Accordingly, it ispossible to prevent the second capacitor from being charged/dischargedthrough the neutral phase input line, which accordingly prevents a lossof symmetry of the input currents.

Further, after the switch is brought into nonconductoin, thecurrent-source power converter converts the multi-phase AC voltageapplied between ones of the input lines into the DC voltage to supply tothe first capacitor and the second capacitor, whereby the inputcapacitor is connected in parallel with the first capacitor and thesecond capacitor. The first capacitor and the second capacitor have beenapplied with the voltage subjected to voltage doubler rectificationuntil then, and thus a voltage between both ends of a pair of the firstcapacitor and the second capacitor is larger than a voltage between bothends of the input capacitor. Accordingly, in a case where the inputcapacitor is connected in parallel with the first capacitor and thesecond capacitor, it is possible to effectively prevent the inrushcurrent from flowing from the input capacitor to the first capacitor andthe second capacitor.

According to the fourth aspect of the direct AC power convertingapparatus of the present invention, the resistor is provided in theneutral phase input line, whereby it is possible to supply the DCcurrent to the first capacitor and the second capacitor through theresistor by using any of the input lines.

According to the fifth aspect of the direct AC power convertingapparatus of the present invention, it is possible to compose a carriercurrent component removing filter for removing a carrier currentcomponent by the reactor and the input capacitor. In addition, theresistor and the reactor are connected in parallel with each other,whereby it is possible to reduce pulsation of a voltage of the inputcapacitor in an initial charging period (transient period).

According to the sixth aspect of the direct AC power convertingapparatus of the present invention, the first capacitor and the secondcapacitor are charged in the state of being connected in series witheach other by the rectifying functions of the first, third and fourthdiodes, and discharged in the state of being connected in parallel witheach other. The first capacitor and the second capacitor are chargedwith a regenerative current from an inductive multi-phase load, anddischarged when exceeding a voltage value which is determined based onthe load power factor on the voltage-source power converting apparatusside. That is, it is possible to secure a discharging path by the firstcapacitor and the second capacitor, and accordingly there can beachieved an operation similar to an operation of a system described inNon-Patent Document 1, though it is a passive circuit.

Further, according to the direct AC power converting apparatus of thesixth aspect, which is according to the direct AC power convertingapparatus of the third aspect according to the second aspect, in thenormal operation in which the current-source power converter converts amulti-phase AC voltage into a DC voltage having two potentials, and thevoltage-source power converter converts the DC voltage into asquare-wave-shape AC voltage, the switch is in nonconduction.Accordingly, a voltage between the input lines is applied to a pair ofthe first capacitor and the second capacitor. Therefore, the voltageserving as a reference for discharging the first capacitor and thesecond capacitor in the state of being connected in parallel with eachother becomes a half value, while it is 1/√{square root over (3)} of themaximum value of the voltage between the input lines in the case wherethe switch is in the conductive state. Accordingly, the waveform of theinput current can be improved.

These and other objects, features, aspects and advantages of the presentinvention will become more apparent from the following detaileddescription of the present invention when taken in conjunction with theaccompanying drawings.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 A conceptual configuration diagram showing an example of a motordriving device according to a first embodiment.

FIG. 2 A flowchart showing an operation of a control section.

FIG. 3 A figure showing a voltage Vrn between an input line ACLr and aneutral phase input line ACLn, a conductive/nonconductive state of atransistor Srp, and a conductive/nonconductive state of a transistorSrn.

FIG. 4 A diagram showing a circuit for describing a mechanism forpreventing inrush current.

FIG. 5 A block diagram of the circuit shown in FIG. 4.

FIG. 6 A figure showing a response of a current flowing through acapacitor shown in FIG. 4.

FIG. 7 A figure showing switching modes.

FIG. 8 A figure showing the voltage Vrn between the input line ACLr andthe neutral phase input line ACLn, conductive/nonconductive states ofthe transistors Srp and Srn, transistors Ssp, Ssn, Stp and Stn.

FIG. 9 A figure showing current vectors.

FIG. 10 A conceptual configuration diagram showing another example ofthe motor driving device according to the first embodiment

FIG. 11 A conceptual configuration diagram showing a motor drivingdevice according to a second embodiment.

FIG. 12 A diagram showing a circuit for describing a mechanism forimproving input characteristics to a capacitor.

FIG. 13 A Bode diagram of the circuit shown in FIG. 12.

FIG. 14 A figure showing a voltage between the input line ACLr and aninput line ACLs, a voltage between both ends of a pair of capacitors Crand Ct, a voltage between both ends of a pair of clamp capacitors Cc1and Cc2, and a voltage between DC power supply lines L1 and L2.

FIG. 15 A graph showing power supply phase voltages Vrn, Vsn and Vtn, aclamp voltage Vc2, power supply line currents Ir, Is and It, and a powersupply neutral point line current In, which are obtained in a case wherea normal operation is performed while a switch S1 is held in conductionin the first embodiment.

FIG. 16 A diagram showing an equivalent circuit of a direct AC powerconverting apparatus according to the first embodiment in an area 1.

FIG. 17 A diagram showing the equivalent circuit of the direct AC powerconverting apparatus according to the first embodiment in an area 2.

FIG. 18 A diagram showing an example of a conceptual configuration of adirect AC power converting apparatus according to a third embodiment.

FIG. 19 A graph showing the power supply phase voltages Vrn, Vsn andVtn, the clamp voltage Vc2, the power supply lines Ir, Is and It, andthe power supply neutral point line current In, which are obtained in acase where a normal operation is performed in the third embodiment.

FIG. 20 A diagram showing an equivalent circuit of the direct AC powerconverting apparatus according to the third embodiment in an area 1.

FIG. 21 A diagram showing the equivalent circuit of the direct AC powerconverting apparatus according to the third embodiment in an area 2.

FIG. 22 A diagram showing another example of the conceptualconfiguration of the direct AC power converting apparatus according tothe third embodiment.

FIG. 23 A graph showing the power supply phase voltages Vrn, Vsn andVtn, and the voltage between the DC power supply lines L1 and L2, whichare obtained in a case where a normal operation is performed in thedirect power converting apparatus according to the third embodiment.

FIG. 24 A configuration diagram showing a power converting apparatus ofNon-Patent Document 1.

FIG. 25 A graph showing a relationship between electrostatic capacitanceof a clamp capacitor and a voltage between both ends of the clampcapacitor.

FIG. 26 A graph showing a relationship between the electrostaticcapacitance of the clamp capacitor and inrush current of the clampcapacitor.

BEST MODE FOR CARRYING OUT THE INVENTION First Embodiment

FIG. 1 shows a conceptual configuration of a motor driving device as anexample of a direct AC power converting apparatus according to a firstembodiment of the present invention. The motor driving device includes apower supply E1, input lines ACLr, ACLs and ACLt, a neutral phase inputline ACLn, reactors Lr, Ls and Lt, capacitors Cr, Cs and Ct, acurrent-source converter 1, DC power supply lines L1 and L2, a clampcircuit 2, a voltage-source inverter 3, a motor 4, a control section 5,a resistor R1 and a switch S1.

The power supply E1 is a multi-phase AC power supply including a neutralpoint (not shown), which is, for example, a three-phase AC power supply.The input lines ACLr, ACLs and ACLt are supplied with an output of thepower supply E1.

The reactors Lr, Ls and Lt are provided on the input lines ACLr, ACLsand ACLt, respectively.

Each of the capacitors Cr, Cs and Ct is connected between ones of theinput lines ACLr, ACLs and ACLt through, for example, Y-connection. Morespecifically, the capacitors Cr and Cs are connected in series betweenthe input lines ACLr and ACLs, the capacitors Cs and Ct are connected inseries between the input lines ACLs and ACLt, and the capacitors Ct andCr are connected in series between the input lines ACLt and ACLr. Thoseare provided on an input side of the current-source converter 1 andfunction as a voltage source. The capacitors Cr, Cs and Ct areconsidered to be input capacitors. On the other hand, the capacitors Cr,Cs and Ct are also considered to constitute, together with the reactorsLr, Ls and Lt, a carrier current component removing filter for removinga carrier current component, respectively.

The current-source converter 1 includes a plurality of switch devices,and converts a three-phase AC voltage applied to ones of the input linesACLr, ACLs and ACLt into a square-wave-shape DC voltage having twopotentials by selection operations of the plurality of switch devices,to thereby supply the DC voltage between the DC power supply lines L1and L2. Note that the DC power supply line L1 is considered to be apositive-side DC power supply line, and that the DC power supply line L2is considered to be a negative-side DC power supply line to which apotential lower than that of the DC power supply line L1 is applied.

More specifically, the current-source converter 1 includes transistorsSrp, Srn, Ssp, Ssn, Stp and Stn, and diodes Drp, Drn, Dsp, Dsn, Dtp andDtn.

Respective cathodes of the diodes Drp, Dsp and Dtp are connected to theDC power supply line L1. Respective anodes of the diodes Drn, Dsn andDtn are connected to the DC power supply line L2.

Emitters of the transistors Srp, Ssp and Stp are connected to anodes ofthe diodes Drp, Dsp and Dtp, respectively. Collectors of the transistorsSrn, Ssn and Stn are connected to cathodes of the diodes Drn, Dsn andDtn, respectively. A collector of the transistor Srp and an emitter ofthe transistor Srn, a collector of the transistor Ssp and an emitter ofthe transistor Ssn, and a collector of the transistor Stp and an emitterof the transistor Stn are connected in common to the input lines ACLr,ACLs and ACLt, respectively.

Respective bases of those transistors Srp, Srn, Ssp, Ssn, Stp and Stnare supplied with a switch signal by the control section 5, and thecurrent-source converter 1 converts the three-phase AC voltage into asquare-wave-shape DC voltage having two potentials.

The clamp circuit 2 includes clamp capacitors Cc1 and Cc2 and a diodeD1. The diode D1 is connected between the DC power supply lines L1 andL2, with an anode and a cathode thereof connected to the DC power supplyline L1 side and the DC power supply line L2 side, respectively. Theclamp capacitors Cc1 and Cc2 are both connected in series with the diodeD1. The clamp capacitors Cc1 and Cc2 and the diode D1 are connected inseries with each other. With the clamp circuit 2 as described above, itis possible to suppress a rise in voltage between the DC power supplylines L1 and L2, which arises from a reflux current flowing from thevoltage-source inverter 3 toward the current-source converter 1. Inaddition, the clamp capacitors Cc1 and Cc2 divide the voltage betweenthe DC power supply lines L1 and L2, and accordingly it is possible toreduce a voltage between both ends of each of the clamp capacitors Cc1and Cc2.

The neutral phase input line ACLn connects the neutral point of thepower supply E1 and a point between the clamp capacitors Cc1 and Cc2.The resistor R1 is inserted in the neutral phase input line ACLn. Theswitch S1 is provided in series with the resistor R1 on the neutralphase input line ACLn.

The voltage-source inverter 3 converts the square-wave-shape DC voltagebetween the DC power supply lines L1 and L2 into a square-wave-shape ACvoltage and outputs to the motor 4. More specifically, thevoltage-source inverter 3 includes transistors Sup, Sun, Svp, Svn, Swpand Swn, and diodes Dup, Dun, Dvp, Dvn, Dwp and Dwn.

Respective collectors of the transistors Sup, Svp and Swp and respectivecathodes of the diodes Dup, Dvp and Dwp are connected to the DC powersupply line L1, and respective emitters of the transistors Sun, Svn andSwn and respective anodes of the diodes Dun, Dvn and Dwn are connectedto the DC power supply line L2.

An emitter of the transistor Sup, a collector of the transistor Sun, ananode of the diode Dup and a cathode of the diode Dun are connected incommon to the motor 4, an emitter of the transistor Svp, a collector ofthe transistor Svn, an anode of the diode Dvp and a cathode of the diodeDvn are connected in common to the motor 4, and an emitter of thetransistor Swp, a collector of the transistor Swn, an anode of the diodeDwp and a cathode of the diode Dwn are connected in common to the motor4.

Bases of those transistors Sup, Sun, Svp, Svn, Swp and Swn are suppliedwith the switch signal by, for example, the control section 5, and thevoltage-source inverter 3 converts the square-wave-shape DC voltagebetween the DC power supply lines L1 and L2 into a square-wave-shape ACvoltage and outputs to the motor 4.

The motor 4 is, for example, a three-phase AC motor, and an inductancecomponent and a resistance component thereof are represented by coilsLu, Lv and Lw, and resistors Ru, Rv and Rw which are connected in seriestherewith, respectively. Those series connections correspond torespective phases of the motor 4. One ends of those series connectionsare connected between the transistors Sup and Sun, between thetransistors Svp and Svn, and between the transistors Swp and Swn,respectively. The other ends of those series connections are connectedin common at a neutral point N.

The voltage-source inverter 3 supplies the square-wave-shape AC voltage.Thanks to the inductance component of the motor 4, an AC current fordriving the motor 4 is smoothed. In other words, the motor 4 convertsthe square-wave-shape AC voltage supplied from the voltage-sourceinverter 3 into the AC current.

The capacitors Cr, Cs and Ct are charged with the AC current flowingthrough the motor 4 via the voltage-source inverter 3 and thecurrent-source converter 1, which is converted into the AC voltage.Becoming reduced, the motor 4 is also considered to be a current sourcefor the current-source converter 1.

The control section 5 controls selection operations of the switch S1 andtransistors of the current-source converter 1. The control section 5controls the selection operation of the transistors of thecurrent-source converter 1 in a sate where the switch S1 is inconduction, and performs voltage doubler rectification on a line currentfor one phase, which flows through one (for example, input line ACLr) ofthe input lines and the neutral phase input line ACLn via the resistorR1 to supply to the clamp capacitors Cc1 and Cc2, to thereby bring theswitch S1 into nonconduction after a lapse of a given period of time.

More specifically, the control section 5 includes an energizationdetecting and synchronization signal generating section 51 and aswitching control section 52.

The energization detecting and synchronization signal generating section51 detects the AC currents flowing through, for example, given twophases (for example, input lines ACLr and ACLs) of the power supply E1to generate a synchronization signal, and supplies the synchronizationsignal to the switching control section 52. In addition, theenergization detecting and synchronization signal generating section 51supplies a switch signal to the switch S1.

The switching control section 52 supplies switching signals to thetransistors of the current-source converter 1 in synchronization withthe input synchronization signal.

An operation of the control section 5 in the motor driving device havingthe configuration as described above will be described. FIG. 2 is aflowchart showing the operation of the control section 5.

First, in Step ST1, the energization detecting and synchronizationsignal generating section 51 receives an activation command from, forexample, an external CPU etc. Then, in Step ST2, the energizationdetecting and synchronization signal generating section 51 which hasreceived the activation command detects the AC currents flowing through,for example, the given two phases (for example, input lines ACLr andACLs) of the power supply E1, and generates a synchronization signalbased on a period of the AC currents, to thereby supply to the switchingcontrol section 52. Accordingly, the energization detecting andsynchronization signal generating section 51 and the switching controlsection 52 are respectively capable of outputting switch signals insynchronization with each other.

Next, in Step ST3, the energization detecting and synchronization signalgenerating section 51 sends the switch signal to the switch 51 to bringthe switch S1 into conduction.

Then, in Step ST4, in synchronization with the received synchronizationsignal, the switching control section 52 performs voltage doublerrectification on a phase voltage for one phase between one (for example,input line ACLr) of the input lines and the neutral phase input lineACLn for charging of the clamp capacitors Cc1 and Cc2. Specifically, theswitching control section 52 starts, for example, the control of thetransistors Srp and Srn. FIG. 3 is a figure showing a voltage Vrnbetween the input line ACLr and the neutral phase input line ACLn,conductive/nonconductive states of the transistors Srp and Srn. Notethat in FIG. 3 the voltage Vrn is positive when the voltage Vrn has highpotential on the input line ACLr side.

As shown in FIG. 3, the switching control section 52 brings thetransistor Srp into conduction in a half period in which the voltage Vrnhas high potential on the input line ACLr side, and brings thetransistor Srn into conduction in the other half period. The clampcapacitor Cc1 is charged in a state where the transistor Srp is inconduction, whereas the clamp capacitor Cc2 is charged in a state wherethe transistor Srn is in conduction. In this case, the resistor R1 isinserted in both charging paths of the clamp capacitors Cc1 and Cc2,whereby the DC voltage is applied to the clamp capacitors Cc1 and Cc2through the resistor R1. Therefore, it is possible to prevent the inrushcurrent from flowing from the power supply E1 to the clamp capacitorsCc1 and Cc2.

The following description is given of a reason why the inrush currentcan be prevented by current flowing through the clamp capacitors Cc1 andCc2 via the resistor R1. For the sake of simplicity, description isgiven of a current i flowing through the circuit in a case where a powersupply voltage Vs (corresponding to the voltage between the input lineACLr and the neutral phase input line ACLn) is applied in series to thecircuit including a reactor L (corresponding to the reactor Lr), aresistor R (corresponding to the resistor R1), and a capacitor C(corresponding to the clamp capacitors Cc1 and Cc2) which are connectedin series with each other.

FIG. 4 is a diagram showing this circuit, and FIG. 5 is a block diagramin which a current ic flowing through the capacitor C when the powersupply voltage Vs is input is assumed to be an output. A transfercharacteristic G(s) of the current ic to the power supply voltage Vs issimilar to Expression (1). When a response to a step input isdetermined, Expression (2) is derived. Here, a resistance value R of theresistor R1 is large, and a transient response (within a range of smalls) is taken into account, whereby the following expression is derived ifthe transfer characteristic is approximated with time-lag of firstorder.

$\begin{matrix}{\lbrack {{Expression}\mspace{14mu} 8} \rbrack \mspace{596mu}} & \; \\{{G(s)} = {\frac{D}{{sE} + F} = \frac{D/E}{s + {F/E}}}} & (8)\end{matrix}$

This is subjected to inverse Laplace transform, whereby the followingexpression is derived.

$\begin{matrix}{\lbrack {{Expression}\mspace{14mu} 9} \rbrack \mspace{596mu}} & \; \\{{{ic}(t)} = {\frac{D}{E}^{{{- F}/E}\; t}}} & (9)\end{matrix}$

Here, D=1/L, E=R/L, and F=1/LC.

FIG. 6 represents Expression (9) graphically, which shows therelationship between the current flowing through the capacitor and time.Note that FIG. 6 shows the results obtained in a case where aninductance of the reactor L is 1 mH, an electrostatic capacitance of thecapacitor C is 330 μF, a resistance value of the resistor R is 10Ω, andthe power supply voltage Vs is 400 V. The maximum value of the currentis obtained by substituting t=0 into Expression (9), and ic(0)=1/R(constant). This is considered to be inrush current, and the inrushcurrent has a value expressed only by the resistance value R. Therefore,it is possible to restrict the inrush current.

Referring to FIG. 2 again, in Step ST5, the energization detecting andsynchronization signal generating section 51 determines whether or not apredetermined period of time has passed from the start of Step ST4, andexecutes Step ST5 again if it has not passed. If it has passed, in StepST6, the energization detecting and synchronization signal generatingsection 51 brings the switch 51 into nonconduction. Since the switch S1is in nonconduction, the AC current from the power supply E1 does notflow through the resistor R1. Accordingly, it is possible to prevent thegeneration of loss produced in the resistor R1 after restricting theinrush current.

Further, in Step ST4, the clamp capacitors Cc1 and Cc2 are supplied withthe phase voltage for one phase between, for example, the input lineACLr and the neutral phase input line ACLn, and thus the capacitors Cr,Cs and Ct are not connected to the clamp capacitors Cc1 and Cc2.Accordingly, it is possible to prevent the inrush current from flowingfrom the capacitors Cr, Cs and Ct to the clamp capacitors even if thecapacitors Cr, Cs and Ct are charged.

Next, in Step ST7, the direct AC power converting apparatus isactivated, to thereby shift to a normal operation. More specifically, inorder to switch the switching operation of the current-source converter1 to a normal operation, the current-source converter 1 is activatedagain, and the voltage-source inverter 3 is also activated. In thenormal operation, the switching control section 52 supplies switchsignals to the transistors Srp, Srn, Ssp, Ssn, Stp and Stn to operatethe current-source converter 1, thereby converting the AC voltage inputfrom the input lines ACLr, ACLs and ACLt into a square-wave-shape DCvoltage having two potentials to supply to the DC power supply lines L1and L2. Then, the voltage-source inverter 3 operates, for example, insynchronization with the current-source converter 1, and converts thesquare-wave-shape DC voltage between the DC power supply lines L1 and L2into a square-wave-shape AC voltage to apply to the motor 4.

The clamp capacitors Cc1 and Cc2 are applied with the DC voltagesubjected to voltage doubler rectification in Step ST4, and thus avoltage between both ends of a pair of the clamp capacitors Cc1 and Cc2is higher than, for example, a voltage between both ends of a pair ofthe capacitors Cr and Cs (specifically, 2/√{square root over (3)}times). Accordingly, it is possible to, in starting the normaloperation, effectively prevent the current initially flowing from thecapacitors Cr and Cs to the clamp capacitors Cc1 and Cc2 from flowing asthe inrush current.

As described above, according to this motor driving apparatus, it ispossible to prevent inrush current from flowing from the power supply E1to the clamp capacitors Cc1 and Cc2. Moreover, it is possible to, instarting the normal operation, effectively prevent inrush current fromflowing from the capacitors Cr, Cs and Ct to the clamp capacitors Cc1and Cc2.

Note that a current-source converter is not typically provided with acurrent limiting resistor because a reactor is typically provided foroutput of a current-source converter. However, in a case where ACvoltage is converted into square-wave-shape voltage having twopotentials and there are provided the clamp capacitors Cc1 and Cc2 whichfunction as capacitors as described above, it is desired to provide acurrent limiting resistor for preventing charging current whichinitially flows from flowing to those as inrush current.

Note that in Step ST4, the switching control section 52 brings thetransistor Srp into conduction in a half period in which the voltage Vrnhas high potential on the input line ACLr side and brings the transistorSrn into conduction in the other half period, which is not limitedthereto. For example, there may be used a switching operation waveformof the current-source converter 1 for one phase in the normal operation.

FIGS. 7 to 9 are figures for describing the switching operation of thecurrent-source converter 1 in the normal operation. In the normaloperation, the switching control section 52 outputs switch signals tothe current-source converter 1 so that six switching modes are selectedin succession as shown in, for example, FIG. 7. Note that in FIG. 7, “1”and “0” represent a state in which the transistor is in conduction and astate in which the transistor is in nonconduction, respectively, andthat I(P) (P is any of rs, rt, st, sr, tr and ts) represents a currentvector output from the current-source converter 1 in accordance with theswitching mode.

FIG. 8 shows the voltage Vrn between the input line ACLr and the neutralphase input line ACLn, and conduction/nonconduction states of thetransistors Srp, Srn, Ssp, Ssn, Stp and Stn. Note that theconduction/nonconduction in the normal operation is indicated by abroken line, and the conductive states of the transistors Srn, Ssn andStn are represented by “−1”. In addition, for example, the transistorsare shifted linearly from the 0 (nonconduction) state to the 1 or −1(conduction) state. If, for example, a pulse width of a switch signalsupplied to a transistor is controlled, the switch operation asdescribed above can be substantially achieved.

The current vector output from the current-source converter 1 describesa locus of a hexagon with respective current vectors I(P) being asvertices by the switching operation shown in FIG. 8, as shown in, forexample, FIG. 9. Through the switching operation as described above, inthe normal operation, the switching control section 52 outputs switchsignals to the transistors Srp, Ssp, Stp, Srn, Ssn and Stn, and convertsa three-phase AC voltage input from the input lines ACLr, ACLs and ACLtinto a square-wave-shape DC voltage having two potentials to supply tothe DC power supply lines L1 and L2.

This operation in a normal operation is applied to the operation in StepST4. More specifically, in Step ST4, the switching control section 52controls the selection operations of the transistors Srp and Srn as, forexample, indicated by a solid line in FIG. 8. This is achieved byoutputting switch signals to the transistors Srp and Srn so that thenearest current vector is output within a phase angle (ωt) of each modeshown in FIG. 9. Note that the selection operation of the transistorsSsp and Ssn may be controlled.

Even in this case, the resistor R1 is arranged in the case where theclamp capacitors Cc1 and Cc2 are charged using the phase voltage for onephase between the input line ACLr and the neutral phase input line ACLn,which prevents the inrush current from flowing from the power supply E1to the clamp capacitors Cc1 and Cc2. In addition, the voltage doublerrectification is performed for charging the clamp capacitors Cc1 andCc2, which also prevents the inrush current from flowing from thecapacitors Cr, Cs and Ct to the clamp capacitors Cc1 and Cc2 in startingthe normal operation.

Further, the switching operation in the normal operation is applicable,whereby there is no need to generate a waveform dedicated to chargingcontrary to, for example, the case shown in FIG. 3.

Further, the switching operation of the current-source converter 1 inStep ST4 and the switching operation of the current-source converter 1in the normal operation can be switched without activating thecurrent-source converter 1 again, whereby it is not necessarily requiredto activate the current-source converter 1 again in Step ST7.

FIG. 10 shows a conceptual configuration of the motor driving device asthe other example of the direct AC power converting apparatus accordingto the first embodiment. The motor driving device shown in FIG. 10 isthe same as the motor driving device shown in FIG. 1 except for theclamp circuit 2. Note that in FIG. 10, circuits at the stages subsequentto the clamp circuit 2 are omitted.

The clamp circuit 2 includes the clamp capacitors Cc1 and Cc2 and diodesD2 to D4. The clamp capacitor Cc1 is connected between the DC powersupply lines L1 and L2. The clamp capacitor Cc2 is connected in serieswith the clamp capacitor Cc1 and is provided on the DC power supply lineL2 side with respect to the clamp capacitor Cc1.

The diode D2 is connected between the clamp capacitors Cc1 and Cc2, withits anode connected to the clamp capacitor Cc1 and its cathode connectedto the clamp capacitor Cc2. The diode D3 has an anode connected betweenthe clamp capacitor Cc2 and the diode D2 and a cathode connected to theDC power supply line L1. The diode D4 has an anode connected to the DCpower supply line L2 and a cathode connected between the clamp capacitorCc1 and the diode D2.

With the clamp circuit 2 as described above, in a case where a currentflowing through the motor 4 delays with respect to a voltage between theDC power supply lines L1 and L2 due to a power load factor on thevoltage-source inverter 3 side, a reflux current flows from the motor 4to the DC power supply lines L1 and L2 in a given period, with theresult that the clamp capacitors Cc1 and Cc2 are charged in a state ofbeing connected in series with each other. A charging voltage (voltagebetween both ends of a pair of the clamp capacitors Cc1 and Cc2) on thisoccasion is determined based on the load power factor as well. On theother hand, the clamp capacitors Cc1 and Cc2 are discharged in a stateof being connected in parallel with each other when respective voltagesbetween both ends of the clamp capacitors Cc1 and Cc2 rise higher than avoltage which is lower one of the square-wave-shape DC voltages betweenthe DC power supply lines L1 and L2. Note that the clamp capacitors Cc1and Cc2 are charged in the state of being connected in series with eachother and discharged in the state of being connected in parallel witheach other, and accordingly a discharging voltage is a half of thecharging voltage.

Through the charging/discharging operation as described above, thevoltages of the clamp capacitors Cc1 and Cc2 are balanced in a casewhere the discharging current is larger than the charging current.

As described above, the reflux current from the motor 4 is charged andis discharged to be supplied to the motor 4 again, with the result thatthe motor 4 is driven with efficiency. In addition, the clamp circuit 2does not require a so-called active device such as a switch device,whereby power consumption and manufacturing cost are reduced.

Second Embodiment

FIG. 11 shows a conceptual configuration of a motor driving device as anexample of a direct AC power converting apparatus according to a secondembodiment. The conceptual configuration of this motor driving device isthe same as the motor driving device shown in FIG. 10 except forresistors R1 to R3 and an auxiliary switch Sr. Note that in FIG. 11,circuits at stages subsequent to the clamp circuit 2 are omitted. Inaddition, the clamp circuit 2 may be one shown in FIG. 1.

The resistors R1 to R3 are inserted in the input lines ACLr, ACLs andACLt, respectively. The auxiliary switch Sr is connected in series withany of the reactors Lr to Lt, and is connected in series with thereactor Lr in FIG. 9. The reactors Ls and Lt are connected in parallelwith the resistors R2 and R3, respectively. A pair of the auxiliaryswitch Sr and the reactor Lr are connected in parallel with the resistorR1.

The energization detecting and synchronization signal generating section51 is capable of controlling a selection operation of the auxiliaryswitch Sr.

The operation of the control section 5 in the above-mentioned motordriving device is the same as that of the flowchart shown in FIG. 2except for Step ST6. In Step ST6, the energization detecting andsynchronization signal generating section 51 brings the switch S1 intononconduction and the auxiliary switch Sr into conduction. Note that inStep ST4, the switching control section 52 desirably outputs switchsignals to the current-source converter 1 as shown in FIG. 3 or asindicated by the solid line of FIG. 8.

Note that the switch S1 is brought into conduction in Step ST3, and theauxiliary switch Sr is in nonconduction at a time when thecurrent-source converter 1 is controlled in Step ST4, whereby the ACcurrents flowing through the input line ACLr and the neutral phase inputline ACLn inevitably flow through the resistor R1. Therefore, it ispossible to effectively prevent the inrush current from flowing into theclamp capacitors Cc1 and Cc2. Note that, without providing the auxiliaryswitch Sr, part of the AC currents flowing through the input line ACLrand the neutral phase input line ACLn may flow into the clamp capacitorsCc1 and Cc2 through the reactor Lr. Even in this case, other part of theAC current flows through the resistor R1, whereby the inrush current canbe reduced. In addition, as to the AC current flowing through thereactor Lr, it is possible to reduce the inrush current thanks to, forexample, a resistance component of the reactor Lr.

Further, in the carrier current component removing filters composed ofthe reactors Lr, Ls and Lt and the capacitors Cr, Cs and Ct,respectively, the resistors R1 to R3 are capable of improvinginput/output transient characteristics of the capacitors Cr, Cs and Ct.This will be described below in detail.

For the sake of simplicity, description is given of a case where thepower supply voltage Vs is input to a circuit in which the capacitor C(corresponding to the capacitors Cr, Cs and Ct) is connected in serieswith a pair of the reactor L (corresponding to the reactors Lr, Ls andLt) and the resistor R (corresponding to the resistors R1 to R3) whichare connected in parallel with each other. FIG. 12 is a configurationdiagram showing this circuit. In this circuit, a voltage V0 between bothends of the capacitor C is considered to be an output in inputting thepower supply voltage Vs. A transfer function of the voltage V0 betweenboth ends to the power supply voltage Vs is as follows.

$\begin{matrix}{\lbrack {{Expression}\mspace{14mu} 10} \rbrack \mspace{571mu}} & \; \\{{G(s)} = {\frac{Vc}{Vs} = {( {{{sL}/R} + 1} )\frac{1/{LC}}{s^{2} + {s/{CR}} + {1/{LC}}}}}} & (10)\end{matrix}$

In this transfer function, undamped natural frequencies f1 and f2 and adamping coefficient ξ are represented by the following expression.

$\begin{matrix}{\lbrack {{Expression}\mspace{14mu} 11} \rbrack \mspace{571mu}} & \; \\{{f_{1} = \frac{1}{2\pi \; {L/R}}},{f_{2} = \frac{1}{2\pi \sqrt{LC}}},{\xi = {\frac{1}{2R}\sqrt{\frac{L}{C}}}}} & (11)\end{matrix}$

FIG. 13 is a Bode diagram showing frequency characteristics of thetransfer function. FIG. 14 shows the results which are obtained in threecases where a resistance value of the resistor R is 10Ω, 30Ω and 100Ω,where an inductance of the reactor L is 1.5 mH and an electrostaticcapacitance of the capacitor C is 10 μF.

FIG. 14 shows a voltage Vrt, a voltage between both ends of a pair ofthe capacitors Cr and Ct, a voltage between both ends of a pair of theclamp capacitors Cc1 and Cc2, and a voltage between the DC power supplylines L1 and L2 in the motor driving device of FIG. 11 using the carriercurrent component removing filter as described above. Note that FIG. 14shows the results when the resistance value of the resistor R1 is 10Ωand 100Ω.

As shown in FIG. 14, damping is produced by the resistance value of theresistor R1, whereby it is possible to reduce voltages (transientvoltages) applied to the capacitors Cr and Ct and the clamp capacitorsCc1 and Cc2 in the transient period (see the results of 10Ω-resistancevalue and 100Ω-resistance value).

Compared with the inrush current shown in FIG. 6, it is possible toreduce the inrush current and also reduce the transient voltages of thecapacitors Cr, Cs and Ct if the resistance value is approximately 10Ω.

Third Embodiment

When the direct AC power converting apparatus described in the firstembodiment is shifted to the normal operation while the switch S1 isheld in conduction due to, for example, a malfunction of the switch S1or control, the power supply line currents flowing through the inputlines ACLr, ACLs and ACLt become asymmetrical, which may cause currentdistortion or overcurrent. First, this problem will be described indetail with reference to the direct AC power converting apparatus shownin FIG. 1.

In the first embodiment, the description has been given assuming thatthe switching operation indicated by the broken line shown in FIG. 8 isperformed in the transistors Srp, Srn, Ssp, Ssn, Stp and Stn in thenormal operation. Here, description is given assuming that forsimplicity of description, the switching operation indicated by thesolid line shown in FIG. 8 is performed in the normal operation.

FIG. 15 shows power supply phase voltages Vrn, Vsn and Vtn, a voltageVc2 between both ends of the clamp capacitor Cc2, power supply linecurrents Ir, Is and It, and a power supply neutral point line currentIn, which are obtained in a case where the normal operation is performedin a state in which the switch S1 is in conduction. Note that the powersupply phase voltages Vrn, Vsn and Vtn have potentials of the inputlines ACLr, ACLs and ACLt with the potential of the neutral phase inputline ACLn being as a reference, respectively. The power supply linecurrents Ir, Is and It are currents flowing through the input linesACLr, ACLs and ACLt, respectively, where a direction of the currentflowing from the power supply E1 to the current-source converter 1 isconsidered to be positive. The power supply neutral point line currentIn is a current flowing through the neutral phase input line ACLn, wherea direction of the current flowing from the current-source converter 1to the power supply E1 is considered to be positive.

In an area 1 shown in FIG. 15, the transistors Srp and Ssn are inconduction (see also FIG. 8). FIG. 16 shows an equivalent circuit of thedirect AC power converting apparatus in the area 1. Note that in FIG.16, a pair of the voltage-source inverter 3 and the motor 4 are shown asa load R0, and the power supply E1 is shown as single-phase powersupplies Er1, Es1 and Et1 which are star-connected. Further, for thesake of simplicity, the current limiting resistor R1 is neglected.

Through the operations of Steps ST1 to ST5 described in the firstembodiment, the clamp capacitors Cc1 and Cc2 are charged with themaximum value V0 of the power supply phase voltages Vrn, Vsn and Vtn.

The transistors Srp and Ssn are brought into conduction in the area 1,whereby a voltage Vrs (=Vrn−Vsn, which is hereinafter referred to as aline voltage Vrs) between the input lines ACLr and ACLs is appliedbetween the DC power supply lines L1 and L2. Accordingly, the currentflows from the single-phase power supply Er1 to the load R0 through thetransistor Srp. In FIG. 16, this is indicated by a power supply linecurrent Ir flowing through the input line ACLr.

On this occasion, the clamp capacitor Cc1 is not charged/discharged. Thereason for this will be described below. The diode D1 impedesdischarging of the clamp capacitor Cc1, and thus a discharging currentdoes not flow from the clamp capacitor Cc1. As a result, a voltage Vc1between both ends of the clamp capacitor Cc1 (hereinafter, referred toas clamp voltage Vc1) is held at the maximum value V0 of the powersupply phase voltage. Accordingly, the clamp voltage Vc1 does not becomesmaller than the power supply phase voltage Vrn, whereby the chargingcurrent does not flow from the DC power supply line L1 to the clampcapacitor Cc1.

On the other hand, the clamp capacitor Cc2 is discharged. The reason forthis will be described below. In the area 1, the current which hasflowed from the input line ACLr to the load R0 initially flows throughthe input line ACLs as it is. On this occasion, the clamp capacitor Cc1and the single-phase power supply Er1 are connected in parallel, andthus the clamp capacitor Cc2 is discharged along with a decrease inabsolute value of the power supply phase voltage Vsn of the single-phasepower supply Es1. The clamp capacitor Cc2 is discharged through theneutral phase input line ACLn. This is indicated by the power supplyneutral point line current In flowing through the neutral phase inputline ACLn in FIGS. 15 and 16.

In the area 1, an absolute value of the power supply line current Ir isequal to a sum of an absolute value of the power supply line current Isand an absolute value of the power supply neutral point line current In.In addition, the power supply line current Ir has a value determined bythe load R0 and the line voltage Vrs, and accordingly, the power supplyline current Is decreases as the absolute value of the power supplyneutral point line current In increases.

A rate of change of the absolute value of the power supply phase voltageVsn increases along with a lapse of time, whereby a rate of drop of theclamp voltage Vc2 also increases along with a lapse of time as well.Therefore, the absolute value of the power supply neutral point linecurrent In increases along with a lapse of time, whereas the absolutevalue of the power supply line current Is decreases in proportion tothis.

Then, the power supply neutral point line current In becomes equal tothe power supply line current Ir when the power supply line current Isbecomes zero. As a result, the clamp voltage Vc2 becomes larger than theabsolute value of the power supply phase voltage Vsn, and thereafter,the power supply neutral point line current In equal to the power supplyline current Ir flows.

As described above, discharging of the clamp capacitor Cc2 through theneutral phase input line ACLn incurs a decrease in the power supply linecurrent Is flowing through the input line ACLs (see FIG. 15).

Next, in an area 2 shown in FIG. 15, the transistors Srp and Stn are inconduction. FIG. 17 shows the equivalent circuit of the direct AC powerconverting apparatus in the area 2.

When the transistors Srp and Stn are brought into conduction in the area2, a voltage Vrt (=Vrn−Vtn, hereinafter also referred to as a linevoltage Vrt) between the input lines ACLr and ACLt is applied betweenthe DC power supply lines L1 and L2. Accordingly, a current flowsthrough the load R0 from the single-phase power supply Er1 via thetransistor Srp.

On this occasion, the clamp capacitor Cc1 is not charged/discharged fromthe same reason as the reason for describing the area 1.

On the other hand, the clamp capacitor Cc2 is mainly charged. The reasonfor this will be described below. With reference to FIG. 15, the clampvoltage Vc2 is smaller than the maximum value V0 of the power supplyphase voltage because of discharging of the clamp capacitor Cc2 in thearea 1. In addition, the clamp voltage Vc2 is initially larger than theabsolute value of the power supply phase voltage Vtn in the area 2.Accordingly, a potential of an anode of the diode Dtn is larger than apotential of a cathode of the diode Dtn, whereby the power supply linecurrent It does not initially flow in the area 2.

Then, the clamp voltage Vc2 also rises as the absolute value of thepower supply phase voltage Vtn increases from a time when the absolutevalue of the power supply phase voltage Vtn exceeds the clamp voltageVc2. In other words, the clamp capacitor Cc2 is charged through theneutral phase input line ACLn. This is indicated by the power supplyneutral point line current In flowing through the neutral phase inputline ACLn in FIGS. 15 and 17.

On this occasion, the current (=power supply line current Ir) flowingthrough the load R0 flows through the input line ACLt via the diode Dtn.Accordingly, the charging current (=absolute value of the power supplyneutral point line current In) of the clamp capacitor Cc2 through theneutral phase input line ACLn and the current through the load R0 flowthrough the input line ACLt. This incurs an increase in the power supplyline current It flowing through the input line ACLt (see FIGS. 15 and17). Note that a rate of change of the absolute value of the powersupply phase voltage Vtn decreases along with a lapse of time, and thusthe charging current flowing into the clamp capacitor Cc2 decreases aswell. Along with this, the absolute value of the power supply linecurrent Is also decreases along with a lapse of time.

As described above, the clamp capacitor Cc2 is repeatedlycharged/discharged through the neutral phase input line ACLn when thenormal operation is performed in the state in which the switch S1 is inconduction, and hence the symmetry of the power supply line currents islost. The direct AC power converting apparatus according to the thirdembodiment is capable of preventing the symmetry of the power supplyline currents as described above from being lost.

FIG. 18 is a diagram showing an example of a conceptual configuration ofthe direct AC power converting apparatus according to the thirdembodiment. Compared with the direct AC power converting apparatus shownin FIG. 1, the switch S1 is not provided in the neutral phase input lineACLn and the clamp circuit 2 further includes a diode D5.

The diode D5 is connected in series with the diode D1 between the DCpower supply lines L1 and L2. An anode and a cathode of the diode D5 arelocated on the DC power supply line L1 side and the DC power supply lineL2 side, respectively. The neutral phase input line ACLn is connectedbetween the diodes D1 and D5.

Charging operations of the clamp capacitors Cc1 and Cc2 are similar tothose of the flowchart shown in FIG. 2. However, since the switch S1 isnot provided, Steps ST3 and ST6 are not required.

FIG. 19 shows the power supply phase voltages Vrn and VsnVtn, the clampvoltage Vc2, the power supply line currents Ir, Is and It, and the powersupply neutral point line current In, which are obtained in a case wherethe normal operation is performed in the direct AC power convertingapparatus shown in FIG. 18.

FIG. 20 shows an equivalent circuit of the direct AC power convertingapparatus according to the third embodiment in the area 1. In a similarmanner as described with reference to FIG. 16, the current flows fromthe single-phase power supply Er1 to the load R0 through the transistorSrp, and the clamp capacitor Cc1 is not charged/discharged on thisoccasion.

On the other hand, the clamp capacitor Cc2 is not charged/discharged aswell. The reason for this will be described below. Discharging of theclamp capacitor Cc2 is impeded by the diode D5. Accordingly, the clampvoltage Vc2 does not fall below the maximum value V0 of the power supplyphase voltage (see the clamp voltage Vc2 of FIG. 19). Therefore, theclamp voltage Vc2 does not fall below the power supply phase voltageVsn, whereby the charging current does not flow from the single-phasepower supply Es1 to the clamp capacitor Cc2 through the neutral phaseinput line ACLn as well.

As described above, the clamp capacitor Cc2 is not charged/discharged,which incurs no increase or decrease in the power supply line currentIs.

FIG. 21 shows the equivalent circuit of the direct AC power convertingapparatus according to the third embodiment in the area 2. In a similarmanner as described with reference to FIG. 17, the current flows fromthe single-phase power supply Er1 to the load R0 through the transistorSrp in the area 1, and the clamp capacitor Cc1 is not charged/dischargedon this occasion.

On the other hand, the clamp capacitor Cc2 is not charged/discharged aswell. The reason for this will be described below. Discharging of theclamp capacitor Cc2 is impeded by the diode D5. Accordingly, the clampvoltage Vc2 is kept at the maximum value V0 of the power supply phasevoltage. Therefore, the clamp voltage Vc2 does not fall below the powersupply phase voltage Vtn, whereby the charging current does not flowfrom the single-phase power supply Et1 to the clamp capacitor Cc2through the neutral phase input line ACLn.

As described above, the clamp capacitor Cc2 is not charged/discharged,which incurs no increase or decrease in the power supply line currentIt.

Accordingly, even if the power supply E1 and the current-sourceconverter 1 are connected to each other through the neutral phase inputline ACLn in the normal operation, it is possible to prevent the clampcapacitor Cc2 from being charged/discharged through the neutral phaseinput line ACLn. This is indicated by the power supply neutral pointline current In in FIG. 19. Therefore, it is possible to prevent a lossof the symmetry of the power supply line currents Ir, Is and It (alsosee the power supply line currents Ir, Is and It of FIG. 19).

FIG. 22 shows another example of the conceptual configuration of thedirect AC power converting apparatus according to the third embodiment.Compared with the direct AC power converting apparatus shown in FIG. 10,the diode D5 is further provided.

The diode D5 is connected, between the DC power supply lines L1 and L2,in series with the diode D5. An anode and a cathode of the diode D5 arelocated on the DC power supply line L1 side and the DC power supply lineL2 side, respectively. The neutral phase input line ACLn is connectedbetween the diodes D2 and D5.

Even when the direct AC power converting apparatus as described above isshifted to the normal operation while the switch S1 is held inconduction, it is possible to prevent the clamp capacitor Cc2 from beingcharged/discharged through the neutral phase input line ACLn, whichprevents a loss of symmetry of the power supply line currents Ir, Is andIt.

Note that while the switch S1 is not necessarily required to be providedon the neutral phase input line ACLn, it is possible to preventdegradation in waveforms of the power supply line currents Ir, Is and Itin the normal operation in a case where the switch S1 is provided toshift to the normal operation with the switch S1 brought intononconcution. This will be described below in detail.

FIG. 23 shows the power supply phase voltages Vrn, Vsn and Vtn, and thevoltage between the DC power supply lines L1 and L2 which are obtainedin a case where the normal operation is performed in this direct ACpower converting apparatus.

As described in the first embodiment, the switching operation indicatedby the broken line in FIG. 8 is performed in the normal operation.Description is given of, for example, a case of a current vector mode 1in which any of the switching modes for outputting the current vectorsI(rs) and I(rt) (also see FIG. 7) is selected. In the current vectormode 1, the transistor Srp is in the conductive state, and thetransistors Ssn and Stn are repeatedly switched therebetween in anexclusive manner. As a result, two line voltages Vrs and Vrt are eachapplied between the DC power supply lines L1 and L2 alternately andrepeatedly. Note that in FIG. 23, peak values of two are shown, and asquare-wave-shape DC voltage between the DC power supply lines L1 andL2, which takes any one of those alternately and repeatedly, is omitted.The description above is also applicable to other current vector mode ifa phase is appropriately read.

Meanwhile, in a case where the normal operation is performed with theswitch S1 brought into nonconduction, as described in the firstembodiment, the reflux current flows through the clamp capacitors Cc1and Cc2 when a load power factor on the voltage-source inverter 3 sidedecreases. For example, in a case where the current flowing through themotor 4 delays with respect to the voltage between the DC power supplylines L1 and L2 due to the load power factor on the voltage-sourceinverter 3 side, the reflux current flows from the motor 4 to the DCpower supply lines L1 and L2 in a given period, whereby the clampcapacitors Cc1 and Cc2 are charged in the state of being connected inseries with each other. The charging voltage (voltage between both endsof a pair of the clamp capacitors Cc1 and Cc2) on this occasion isdetermined based on the load power factor. On the other hand, the clampcapacitors Cc1 and Cc2 are discharged in the state of being connected inparallel with each other when the square-wave-shape DC voltage betweenthe DC power supply lines L1 and L2 falls below the clamp voltages Vc1and Vc2. Note that the clamp capacitors Cc1 and Cc2 are charged in thestate of being connected in series with each other and discharged in thestate of being connected in parallel with each other, and accordinglythe discharging voltage is a half of the charging voltage.

In other words, since the reflux current does not flow in a case wherethe load power factor is one, the clamp capacitors Cc1 and Cc2 are notcharged/discharged. The reason for this will be described below. In thecase where the switch S1 is in nonconduction, the voltages (linevoltages) between ones of the input lines ACLr, ACLs and ACLt areapplied between the power supply lines L1 and L2. Therefore, the voltagebetween both ends of a pair of the clamp capacitors Cc1 and Cc2 is equalto the maximum value V1 of the line voltage. Assuming that theelectrostatic capacitances of the clamp capacitors Cc1 and Cc2 are equalto each other, the clamp voltages Vc1 and Vc2 are a half value of themaximum value V1. On the other hand, the minimum value of thesquare-wave-shape DC voltage between the DC power supply lines L1 and L2is a half value of the maximum value V1 as well. Therefore, the DCvoltage between the DC power supply lines L1 and L2 does not fall belowthe clamp voltages Vc1 and Vc2, whereby the clamp capacitors Cc1 and Cc2are not discharged.

The above description has been given of the operation of the clampcircuit 2 when the normal operation is performed with the switch S1brought into nonconduction and when the load power factor is one. Thatis, the clamp capacitors Cc1 and Cc2 serve a function of causing thereflux current from the motor 4 to flow when the load power factor issmall, and does not contribute to the operation when the load powerfactor is one.

On the other hand, in the case where the normal operation is performedwith the switch S1 being held in conduction, the power supply phasevoltage is applied to each of the clamp capacitors Cc1 and Cc2. Thepower supply phase voltage is 1/√{square root over (3)} of the linevoltage. Therefore, the clamp voltages Vc1 and Vc2 are the maximum valueV0 of the power supply phase voltage, that is, 1/√{square root over (3)}of the maximum value V1 of the line voltage.

The minimum value of the DC voltage between the DC power supply lines L1and L2 is a half value of the maximum value V1, and thus the DC voltagebetween the DC power supply lines L1 and L2 falls below the clampvoltages Vc1 and Vc2 even if the load power factor is one (see FIG. 23).Accordingly, the clamp capacitors Cc1 and Cc2 are discharged in thestate of being connected in parallel with each other in a period inwhich the DC voltage between the DC power supply lines L1 and L2 fallsbelow the clamp voltages Vc1 and Vc2. In this period, the clampcapacitors Cc1 and Cc2 supply an operating current to the motor 4 andthe current from the power supply E1 does not flow through the motor 4.Therefore, the current does not flow through the input lines ACLr, ACLsand ACLt in this period, which degrades a waveform of the power supplyline current.

As described above, when the switch S1 is provided on the neutral phaseinput line ACLn to shift to the normal operation with this brought intononconduction, that is, when Step ST6 of FIG. 2 is performed to shift tothe normal operation, it is possible to prevent the waveform of thepower supply line current from degrading.

Note that the direct AC power converting apparatus according to thethird embodiment may be applied to the direct AC power convertingapparatus according to the second embodiment.

While the invention has been shown and described in detail, theforegoing description is in all aspects illustrative and notrestrictive. It is therefore understood that numerous modifications andvariations can be devised without departing from the scope of theinvention.

1. A direct AC power converting apparatus, comprising: a plurality ofinput lines to which an output of a multi-phase AC power supplyincluding a neutral point is applied; a positive-side DC power supplyline; a negative-side DC power supply line to which a potential lowerthan a potential applied to said positive-side DC power supply line isapplied; a current-source power converter including a plurality ofswitch devices, converting a multi-phase AC voltage applied between onesof said plurality of input lines into a square-wave-shape DC voltagehaving two potentials by selection operations of said plurality ofswitch devices, and supplying said DC voltage between said positive-sideDC power supply line and said negative-side DC power supply line; aplurality of input capacitors each provided between the ones of saidplurality of input lines and functioning as a voltage source; a firstdiode provided between said positive-side DC power supply line and saidnegative-side DC power supply line and having an anode on saidpositive-side DC power supply line and a cathode on said negative-sideDC power supply line side; first and second capacitors connected,between said positive-side DC power supply line and said negative-sideDC power supply line, in series with said first diode; a neutral phaseinput line connecting said neutral point and a point between said firstcapacitor and said second capacitor; a voltage-source power converterconverting said DC voltage into a square-wave-shape AC voltage andoutputting to an inductive multi-phase load; a resistor inserted in anyone of said plurality of input lines and said neutral phase input line;and a control section controlling the selection operations of saidplurality of switch devices, performing voltage doubler rectification ona phase voltage for one phase which is applied between one of saidplurality of input lines and said neutral phase input line, andsupplying for charging of said first capacitor and said second capacitorthrough said resistor.
 2. The direct AC power converting apparatusaccording to claim 1, further comprising: a second diode connected,between said positive-side DC power supply line and said negative-sideDC power supply line, in series with said first diode, and having ananode on said positive-side DC power supply line side and a cathode onsaid negative-side DC power supply line side, wherein said neutral phaseinput line is connected between said first and second diodes.
 3. Thedirect AC power converting apparatus according to claim 1, furthercomprising: a switch provided on said neutral phase input line, whereinsaid control section supplies for charging of said first capacitor andsaid second capacitor in a state in which said switch is in conduction,and brings said switch into nonconduction after a lapse of apredetermined period of time.
 4. The direct AC power convertingapparatus according to claim 2, further comprising: a switch provided onsaid neutral phase input line, wherein said control section supplies forcharging of said first capacitor and said second capacitor in a sate inwhich said switch is in conduction, and brings said switch intononconduction after a lapse of a predetermined period of time.
 5. Thedirect AC power converting apparatus according to claim 1, wherein saidresistor is provided on said neutral phase input line.
 6. The direct ACpower converting apparatus according to claim 2, wherein said resistoris provided on said neutral phase input line.
 7. The direct AC powerconverting apparatus according to claim 3, wherein said resistor isprovided on said neutral phase input line.
 8. The direct AC powerconverting apparatus according to claim 4, wherein said resistor isprovided on said neutral phase input line.
 9. The direct AC powerconverting apparatus according to claim 1, wherein said resistor isprovided in one of said plurality of input lines, the direct AC powerconverting apparatus further comprising reactors connected in parallelwith said resistor.
 10. The direct AC power converting apparatusaccording to claim 2, wherein said resistor is provided in one of saidplurality of input lines, the direct AC power converting apparatusfurther comprising reactors connected in parallel with said resistor.11. The direct AC power converting apparatus according to claim 3,wherein said resistor is provided in one of said plurality of inputlines, the direct AC power converting apparatus further comprisingreactors connected in parallel with said resistor.
 12. The direct ACpower converting apparatus according to claim 4, wherein said resistoris provided in one of said plurality of input lines, the direct AC powerconverting apparatus further comprising reactors connected in parallelwith said resistor.
 13. The direct AC power converting apparatusaccording to claim 1, wherein said first capacitor is provided on saidpositive-side DC power supply line side with respect to said secondcapacitor, and said first diode is provided between said first capacitorand said second capacitor, the direct AC power converting apparatusfurther comprising: a third diode having an anode connected between saidfirst diode and said second capacitor and a cathode connected to saidpositive-side DC power supply line; and a fourth diode having an anodeconnected to said negative-side DC power supply line and a cathodeconnected between said first diode and said first capacitor.